AN2761
Application note
Solution for designing a transition mode
PFC preregulator with the L6562A
Introduction
The TM (transition mode) technique is widely used for power factor correction in low and
middle power applications, such as lamp ballasts, high-end adapters, flat TVs and monitors,
基因
测序仪
and PC power supplies. The L6562A is the latest proposal from STMicroelectronics for this
market as well as emerging markets that may require a low-cost power factor correction.
August 2008 Rev 11/36
www.st
Contents AN2761
Contents
1Introduction to power factor correction . . . . . . . . . . . . . . . . . . . . . . . . . 4 2TM PFC operation (boost topology) . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
3Designing a TM PFC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
3.1Input specification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
3.2Operating condition . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
3.3Power section design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
3.3.1Bridge rectifier . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
3.3.2Input capacitor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
3.3.3Output capacitor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
3.3.4Boost inductor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
3.3.5Power MOSFET selection and dissipation . . . . . . . . . . . . . . . . . . . . . . . 14
3.3.6Boost diode selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
3.4L6562A biasing circuitry . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 4Design example using the L6562A-TM PFC Excel spreadsheet . . . . . 25 5EVL6562A-TM-80W evaluation board . . . . . . . . . . . . . . . . . . . . . . . . . . 27 6Test results and significant waveforms . . . . . . . . . . . . . . . . . . . . . . . . 29 7L6562A layout hints . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33 8Reference . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34 9Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35
2/36
AN2761List of figures List of figures
Figure 1.L6562A PFC controller in an SMPS architecture . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 Figure 2.Boost converter circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 Figure 3.Inductor current waveform and MOSFET timing. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 Figure 4.Switching frequency fixing the line voltage. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14 Figure 5.Transition angle versus input voltage. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16 Figure 6.Capacitive losses . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
Figure 7.Conduction losses and total losses in the STP8NM50 MOSFET for the 80W TM PFC. . . 17 Figure 8.L6562A internal schematic. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 Figure 9.Bode plot - open-loop transfer function. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20 Figure 10.Bode plot - phase . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20 Figure 11.Multiplier characteristics family. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22 Figure 12.Optimum MOSFET turn-on. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24 Figure 13.Excel spreadsheet design specification input table . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25 Figure 14.Other design data. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25 Figure 15.Excel spreadsheet TM PFC schematic. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26 Figure 16.Excel spreadsheet BOM - 80 W TM PFC based on L6562A . . . . . . . . . . . . . . . . . . . . . . . 26 Figure 17.EVL6562A-TM-80W evaluation board . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27 Figure 18.Wide range 80W evaluation board electrical circuit (EVL6562A-TM-80W) . . . . . . . . . . . . 27 Figure 19.EVL6562A-TM-80W compliance to EN61000-3-2 standard. . . . . . . . . . . . . . . . . . . . . . . . 29 Figure 20.EVL6562A-TM-80W compliance to JEIDA-MITI standard . . . . . . . . . . . . . . . . . . . . . . . . . 29 Figure 21.EVL6562A-TM-80W power factor vs. Vin & load . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30 Figure 22.EVL6562A-TM-80W THD vs. Vin & load . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30 Figure 23.EVL6562A-TM-80W efficiency vs. Vin & load. . . . . . . . . . . . . . . . . . . . . . .
. . . . . . . . . . . . 30 Figure 24.EVL6562A-TM-80W static Vout regulation vs. Vin & load. . . . . . . . . . . . . . . . . . . . . . . . . 30 Figure 25.EVL6562A-TM-80W input current at 100 V-50 Hz - 80 W load . . . . . . . . . . . . . . . . . . . . . 31 Figure 26.EVL6562A-TM-80W input current at 230 V-50 Hz - 80 W load . . . . . . . . . . . . . . . . . . . . . 31 Figure 27.EVL6562A-TM-80W input current at 100 V-50 Hz - 40 W load . . . . . . . . . . . . . . . . . . . . . 31 Figure 28.EVL6562A-TM-80W input current at 230 V-50 Hz - 40 W load . . . . . . . . . . . . . . . . . . . . . 31 Figure 29.EVL6562A-TM-80W input current at 100 V-50 Hz - 20 W load . . . . . . . . . . . . . . . . . . . . . 32 Figure 30.EVL6562A-TM-80W input current at 230 V-50 Hz - 20 W load . . . . . . . . . . . . . . . . . . . . . 32
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Introduction to power factor correction AN27614/36
1 Introduction to power factor correction
The front-end stage of conventional offline converters, typically consisting of a full-wave
rectifier bridge with a capacitor filter, has an unregulated DC bus from the AC mains. The
filter capacitor must be large enough to have a relatively low ripple superimposed on the DC
level. This means that the instantaneous line voltage is below the voltage on the capacitor
most of the time, thus the rectifiers conduct only for a small portion of each line half-cycle.
The current drawn from the mains is then a series of narrow pulses whose amplitude is 5-10
times higher than the resulting DC value. Many drawbacks result such as a much higher
分合闸
电磁铁peak and RMS current down from the line, distortion of the AC line voltage, overcurrents inthe neutral line of the three-phase systems and, consequently, a poor utilization of the power
system's energy capability. This can be measured in terms of either total harmonic distortion
管道渗漏检测
(THD), as norms provide for, or power factor (PF), intended as the ratio between the real power (the one transferred to the output) and the apparent power (RMS line voltage times
RMS line current) drawn from the mains, which is more immediate. A traditional input stage
with capacitive filter has a low PF (0.5-0.7) and a high THD (>100%). By using switching
techniques, a power factor corrector (PFC) preregulator, located between the rectifier bridge
and the filter capacitor, allows drawing a quasi-sinusoidal current from the mains, in phase
with the line voltage. The PF becomes very close to 1 (more than 0.99 is possible) and the
previously mentioned drawbacks are eliminated. Theoretically, any switching topology can
be used to achieve a high PF but, in practice, the boost topology has become the most
popular thanks to the advantages it offers:
●
primarily because the circuit requires the fewest external parts (low-cost solution)●the boost inductor located between the bridge and the switch causes the input di/dt to be low, thus minimizing the noise generated at the input and, therefore, the
requirements on the input EMI filter
●
the switch is source-grounded, therefore easy to drive However, boost topology requires the DC output voltage to be higher than the maximum
expected line peak voltage (400 VDC is a typical value for 230 V or wide-range mains
applications). In addition, there is no isolation between the input and output, thus any line
voltage surge is passed on to the output. Two methods of controlling a PFC preregulator are
currently widely used: the fixed frequency average current mode PWM (FF PWM) and the
transition mode (TM) PWM (fixed ON-time, variable frequency). The first method needs a
complex control that requires a sophisticated controller IC (ST's L4981A, with the variant of
the frequency modulation offered by the L4981B) and a considerable component count. The
second one requires a simpler control (implemented by ST's L6562A), much fewer external
parts and is therefore much less expensive. With the first method the boost inductor works
in continuous conduction mode, while TM makes the inductor work on the boundary
between continuous and discontinuous mode, by definition. For a given throughput power,
TM operation involves higher peak currents. This, also consistently with cost considerations,
suggests its use in a lower power range (typically below 200 W), while the former is
recommended for higher power levels. For completion, FF PWM is not the only alternative
when CCM operation is desired. FF PWM modulates both switch ON and OFF times (their
sum is constant by definition), and a given converter operates in either CCM or DCM
depending on the input voltage and the load conditions. Exactly the same result can be
achieved if the ON-time only is modulated and the OFF-time is kept constant, in which case,
however, the switching frequency is no longer fixed. This is referred to as "fixed-OFF-time"
(FOT) control. Peak-current-mode control can still be used. In this application note transition
mode is studied in depth.
AN2761TM PFC operation (boost topology) 5/362 TM PFC operation (boost topology)
The operation of the PFC transition mode controlled boost converter, can be summarized in
the following description.
The AC mains voltage is rectified by a bridge and the rectified voltage is delivered to the
boost converter. This, using a switching technique, boosts the rectified input voltage to a
regulated DC output voltage (Vo).
The boost converter consists of a boost inductor (L), a controlled power switch (Q), a catch
diode (D), an output capacitor (Co) and, obviously, a control circuitry (see Figure 2).
The goal is to shape the input current in a sinusoidal fashion, in phase with the input
瓶嘴sinusoidal voltage. To do this the L6562A uses the transition mode technique.
Figure 2.Boost converter circuit
The error amplifier compares a partition of the output voltage of the boost converter with an
internal reference, generating an error signal proportional to the difference between them. If
the bandwidth of the error amplifier is narrow enough (below 20 Hz), the error signal is a DC
value over a given half-cycle.
laflofThe error signal is fed into the multiplier block and multiplied by a partition of the rectified
mains voltage. The result is a rectified sinusoid whose peak amplitude depends on the
mains peak voltage and the value of the error signal.
The output of the multiplier is in turn fed into the (+) input of the current comparator, thus it
represents a sinusoidal reference for PWM. In fact, as the voltage on the current sense pin
(instantaneous inductor current times the sense resistor) equals the value on the (+) of the
current comparator, the conduction of the MOSFET is terminated. As a consequence, the
电脑控制板peak inductor current is enveloped by a rectified sinusoid. As demonstrated in Section 3.3.4, TM control causes a constant ON-time operation over each line half-cycle.
After the MOSFET has been turned off, the boost inductor discharges its energy into the
load until its current goes to zero. The boost inductor has now run out of energy, the drain
node is floating and the inductor resonates with the total capacitance of the drain. The drain
voltage drops rapidly below the instantaneous line voltage and the signal on ZCD drives the
MOSFET on again and another conversion cycle starts.
This low voltage across the MOSFET at turn-on reduces both the switching losses and the
total drain capacitance energy that is dissipated inside the MOSFET.
The resulting inductor current and the timing intervals of the MOSFET are shown in
Figure 3, where it is also shown that, by geometric relationships, the average input current
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